1. Field of the Invention
The present invention relates to a method of estimating a transmission or telecommunications channel which uses complementary sequences. The method results either in obtaining an optimal estimation of the phase and of the attenuation in the case of a single-path channel if the arrival time of the signal is known, or in obtaining a very effective estimation of the delays, phases and attenuations of the different paths in the case of a multipath channel. The method also makes it possible to obtain an estimation in the case of a channel of which it is not possible to distinguish the different paths or in the case of a multipath channel, of which one of the paths is very powerful in comparison with all the others, as long as the arrival time of the signal is known.
2. Background of the Invention
In a telecommunications system, information circulates between transmitters and receivers through-channels. In this connection, FIG. 1 illustrates a model, which is discrete in time, of the transmission chain between a transmitter 1 and a receiver 2 through a transmission channel 3. As a general rule, the transmission channels can correspond to different physical, radio, wire, optical media etc., and to different environments, fixed or mobile communications, satellites, submarine cables, etc.
As a result of the multiple reflections of which the waves emitted by transmitter 1 can be the object, channel 3 is a multipath channel which is generally modelled as FIG. 1 indicates. It is then considered to be a shift register 30 comprising L serial cells (referred to by a subscript k able to take values of between 1 and L) and the contents of which are shifted towards the right of FIG. 1 each time a symbol arrives at its input. The output of each cell with the subscript k is applied to a filter 31 representing the interference undergone by this output and introducing an attenuation of the amplitude ak, a phase shift xcex1k and a delay rk. The outputs of the filters are summed in a summer 32. The total impulse response thus obtained is marked h(n).
The output of summer 32 is applied to the input of an adder 33 which receives, moreover, a random signal, modelled by a Gaussian white noise, w(n) which corresponds to the thermal noise which is present in the telecommunications system.
In FIG. 1, the reference h(n) has been used, in channel 3, for the register 30, the filters 31 and the summer 32, followed by an adder which adds the noise w(n).
It will be understood that, if the transmitter 1 transmits the signal e(n), the signal received r(n), in the receiver 2, is thus:                               r          ⁡                      (            n            )                          =                                            e              ⁡                              (                n                )                                      *                          h              ⁡                              (                n                )                                              +                      w            ⁡                          (              n              )                                                              =                                            e              ⁡                              (                n                )                                      *                                          ∑                                  k                  =                  1                                L                            ⁢                                                a                  k                                ⁢                                  δ                  ⁡                                      (                                          n                      -                                              r                        k                                                              )                                                  ⁢                                  ⅇ                                      j                    ⁢                                          xe2x80x83                                        ⁢                                          α                      k                                                                                                    +                      w            ⁡                          (              n              )                                                              =                                            ∑                              k                =                1                            L                        ⁢                                          a                k                            ⁢                              e                ⁡                                  (                                      n                    -                                          r                      k                                                        )                                            ⁢                              ⅇ                                  j                  ⁢                                      xe2x80x83                                    ⁢                                      α                    k                                                                                +                      w            ⁡                          (              n              )                                          
In these expressions       h    ⁢          (      n      )        =            ∑              k        =        1            L        ⁢                  a        k            ⁢              δ        ⁡                  (                      n            -                          r              k                                )                    ⁢              ⅇ                  jα          k                    
denotes the impulse response of the channel, xcex4(n) being the Dirac impulse. The operator * denotes the convolution product, defined by the following relation:       c    ⁢          (      n      )        =                    a        ⁢                  (          n          )                    *              b        ⁢                  (          n          )                      =                  ∑                  m          =                      -            ∞                                    +          ∞                    ⁢                        a          ⁢                      (            m            )                          ·                  b          ⁢                      (                          n              -              m                        )                              
Thus it is generally necessary to determine the characteristics of channel 3, at a given moment, in order to thwart the induced distortion of the transmitted signal e(n). In order to obtain an estimation of h(n), i.e. of the coefficients ak, rk and xcex1k of the model of channel 3, it is necessary to repeat this operation at a greater or lesser frequency depending on the rate at which the characteristics of the channel evolve.
A widespread method of estimating the channel consists in transmitting, via transmitter 1, signals e(n) which are predetermined and known to receiver 2, and in comparing the signals received r(n) in receiver 2, by means of a periodic or a periodic correlation, with those which are expected there in order to deduce from them the characteristics of the channel. The a periodic correlation of two signals of length N has a total length 2Nxe2x88x921 and is expressed, from the convolution product, by the relation:                     ϕ                  a          ,          b                    ⁡              (        n        )              =                                        a            *                    ⁡                      (                          -              n                        )                          *                  b          ⁢                      (            n            )                              =                        ∑                      m            =            0                                N            -            1                          ⁢                              a            ⁡                          (              m              )                                ·                                    b              ⁡                              (                                  m                  +                  n                                )                                                    (              1              )                                            ,      
    ⁢            [      m      ]        =    0    ,  1  ,  …  ⁢      xe2x80x83    ,      N    -    1  
for two signals a(n) and h(n) of finite length N, where the operator * denotes the complex conjugate operation.
The correlation of the received signal r(n) with the known transmitted signal e(n) translates as:
r(n)*e*(xe2x88x92n)=[e(n)*h(n)+w(n)]*e*(xe2x88x92n)
"PHgr"e,r(n)="PHgr"e,e*h(n)+"PHgr"e,w(n)
="PHgr"e,e(n)*h(n)+"PHgr"e,w(n)
The result of the correlation operation constitutes the estimation of the impulse response of the channel: the quality or the precision of the estimation is all the better if e(n) tends towards h(n). The latter is directly dependant on the choice of transmitted sequence e(n); to optimise the estimation process, the signal e(n) should be chosen in such a way that "PHgr"e,e(n) tends towards kxc2x7xcex4(n), k being a real number, and that "PHgr"e,w(n)/"PHgr"e,e(n) tends towards zero. In fact, in this case, the estimation of the channel becomes:
"PHgr"e,r(n)=kxc2x7xcex4(n)*h(n)+"PHgr"e,w(n)
=kxc2x7h(n)+"PHgr"e,w(n)
"PHgr"e,r(n)≈kxc2x7h(n)
It has been demonstrated that no single sequence exists for which the function of a periodic auto-correlation "PHgr"e,e(n) tends toward kxc2x7xcex4(n).
One object of the present invention consists in using pairs of complementary sequences which have the property that the sum of their auto-correlations is a perfect Dirac function. Let s(n) and g(n), n=0,1, . . . , Nxe2x88x921 be a pair of complementary sequences:
"PHgr"s,s(n)+"PHgr"g,g(n)=k.xcex4(n)xe2x80x83xe2x80x83(1)
Several methods of constructing such complementary sequences are known in the literature: Golay complementary sequences, polyphase complementary sequences, Welti sequences, etc. By way of information, one will be able to refer, in this connection, to the following technical documents which deal with the introduction to complementary sequences and, in particular, to Golay complementary sequences as well as to a Golay correlator:
1) xe2x80x9cOn a periodic and periodic complementary sequencesxe2x80x9d by Feng K., Shiue P. J. -S., and Xiang Q., published in the technical journal IEEE Transactions on Information Theory, Vol. 45, no. 1, January 1999,
2) xe2x80x9cKorrelationssignalexe2x80x9d by Lxc3xcke H. -D, published in the technical journal ISBN 3-540-54579-4, Springer-Verlag Heidelberg New York, 1992,
3) xe2x80x9cPolypbase Complementary Codesxe2x80x9d by R. L. Frank, published in the technical journal IEEE Transactions on Information Theory, November 1980, Vol. IT26, no. 6,
4) xe2x80x9cMultiphase Complementary Codesxe2x80x9d by R. Sivaswamy, published in the technical journal IEEE Transactions on Information Theory, September 1978, Vol. IT-24, no. 5,
5) xe2x80x9cEfficient pulse compressor for Golay complementary sequencesxe2x80x9d by S. Z. Budissin, published in the technical journal Electronics Letters, Vol. 27, no. 3, January 1991,
6) xe2x80x9cComplementary Seriesxe2x80x9d by M. J. Golay, published in the technical journal IRE Trans; on Information Theoryxe2x80x9dVol. IT-7, April 1961,
7) xe2x80x9cEfficient Golay Correlatorxe2x80x9d by B. M. Popovic, published in the technical journal IEEE Electronics Letters, Vol. 35, no. 17, August 1999.
Reference can also be made to the descriptions of the documents U.S. Pat. Nos. 3,800,248, 4,743,753, 4,968,880, 5,729,612, 5,841,813, 5,862,182 and 5,961,463.
The property of complementary sequences in having a perfect sum of autocorrelations is illustrated in FIG. 2, taking, by way of example, a pair of Golay complementary sequences of length N=16 bits.
In FIG. 2 are plotted on the x-co-ordinates the time shifts in relation to perfect synchronisation. The possible shifts are numbered from 1 to 31 for the pair of sequences s(n) and g(n), and on the y-co-ordinates the correlations from xe2x88x925 to +35. The curve in dashes corresponds to the auto-correlation "PHgr"s,s(n) of the sequence s(n); the curve in a dot-dash line to the auto-correlation "PHgr"g,g(n) of the sequence g(n): and the curve in an unbroken line to the sum of the auto-correlations "PHgr"s,s(n) and "PHgr"g,g(n). One can see that the curve in an unbroken line merges with the axis of the x-co-ordinates between points 0 and 15 and points 17 and 31, but it corresponds practically to a Dirac function between points 15 to 17.
The theoretically perfect auto-correlation properties of these complementary sequences may, however, only be exploited if their transmission can be ensured in such a manner that the occurrence of inter-correlations "PHgr"s,g(n) and /or "PHgr"g,s(n) is avoided.
According to one feature of the invention, a method is provided of estimating a transmission or telecommunications channel, in which method a composite signal of complementary sequences is used and in which a pair of complementary sequences s(n) and g(n) is transmitted after having multiplexed them in phase.
According to another feature of the invention, a method is provided of constructing the composite signal from a pair of polyphase complementary sequences s(n) and g(n) which are multiplexed in phase, this method making it possible to exploit the property "PHgr"s,s(n)+"PHgr"g,g(n) mentioned in the relation (1) above.
According to another feature, the composite signal is made up of two polyphase complementary sequences s(n) and (g(n) transmitted with a phase shift between them of 90xc2x0, i.e. the transmitted composite signal e(n) is in the form of the relation (2) below:
e(n)=ei"PHgr".(s(n)+j.g(n))xe2x80x83xe2x80x83(2)
with an initial, fixed and known phase shift "PHgr".
In the case of binary complementary sequences s(n) and g(n), with a number of phases P equal to 2, i.e. the case of Golay complementary sequences, the transmitted signal e(n) is in the form of a signal 2P-PSK, or 4-PSK, as FIG. 3 shows in the complex plane. FIG. 3 represents, id the complex plane (R,I), the transmitted composite signal e(n), of which the values 0 or 1 taken by each component s(n), g(n) are respectively represented by the ends of a corresponding segment S and G. Segments S and G are out of phase with one another by II/2.
In the more general case of polyphase complementary sequences with a number of phases P greater than 2, the transmitted signal e(n) takes the form of a signal (2P)-PSK.
According to another feature, a device is provided which is intended to generate the composite signal according to relation (2) and which comprises a first generator capable of generating the first sequence s(n), with n varying from 0 to Nxe2x88x921, the output of which is connected to the first input of an adder, and a second generator capable of generating the second sequence g(n), with n varying from 0 to Nxe2x88x921, the output of which is connected to the input of a first circuit shifting phase by 90xc2x0, the output of which is connected to the second input of the adder, the output of the adder being connected to the input of a second circuit shifting phase by "PHgr" which delivers the composite signal.
The features of the present invention mentioned above, as well as others, will appear more clearly in reading the description of embodiments, said description being made in connection with the attached drawings, amongst which: